整流器外文翻譯---無交流電動勢、電流傳感器的三相pwm整流器控制

1、<p>　　AC Voltage and Current Sensorless Control of</p><p>　　Three-Phase PWM Rectifiers</p><p>　　Dong-Choon Lee, Member, IEEE, and Dae-Sik Lim</p><p>　　1 THREE-PHASE PWM RECTIFI

2、ERS</p><p>　　A System Modeling</p><p>　　Fig. 1 shows the power circuit of the three-phase PWM rectifier. The voltage equations are given by</p><p><b>　　(1)</b></p>

3、<p>　　Fig. 1. Three-phase PWM rectifier without ac-side sensors.</p><p>　　where , and are the source voltage, the line current, and the rectifier input voltage, respectively and are the input resistan

4、ce and the input inductance, respectively. When the peak line voltage , angular frequency , and initial phase angle are given, assuming a balanced three-phase system, the source phase voltage is expressed as</p>&

5、lt;p><b>　　(2)</b></p><p><b>　　Where</b></p><p><b>　　(3)</b></p><p>　　A transformation matrix based on the estimated phase angle ,which transfo

6、rms three-phase variables into a synchronous d–q reference frame, is</p><p><b>　　(4)</b></p><p>　　Transforming (1) into the – reference frame using (4)</p><p><b>

7、　　(5)</b></p><p>　　where p is a differential operator and . </p><p>　　Expressing (5) in a vector notation</p><p><b>　　(6)</b></p><p><b>　　where,

8、</b></p><p>　　,,, (7)</p><p>　　Taking a transformation of (2) by using (4)</p><p><b>　　(8)</b></p><p><b>　　Where</b><

9、/p><p><b>　　(9)</b></p><p>　　Expressing (6) and (8) in a discrete domain, by approximating the derivative term in (6) by a forward difference [9], respectively, </p><p><

10、;b>　　(10)</b></p><p><b>　　(11) </b></p><p>　　Where T is the sampling period. </p><p>　　Fig. 2. Overall control block diagram.</p><p>　　B System Con

11、trol</p><p>　　The PI controllers are used to regulate the dc output voltage and the ac input current. For decoupling current control, the cross-coupling terms are compensated in a feed forward-type</p>

12、<p>　　and the source voltage is also compensated as a disturbance. For transient responses without overshoot, the anti-windup technique is employed [10]. The overall control block diagram eliminating the source vo

13、ltage and line current sensors is shown in Fig. 2. The estimation algorithm of source voltages and line currents is described in the following sections.</p><p>　　2 PREDICTIVE CURRENT ESTIMATION</p>&l

14、t;p>　　The currents of and can not be calculated instantly since the calculation time of the DSP is required. To eliminate the delay effect, a state observer can be used. In addition, the state observer provides the

15、filtering effects for the estimated variable.</p><p>　　Expressing (5) in a state-space form,</p><p><b>　　(12)</b></p><p><b>　　(13)</b></p><p>&l

16、t;b>　　where, </b></p><p><b>　　,,</b></p><p><b>　　,</b></p><p>　　And y is the output. </p><p>　　Transforming (12) and (13) into a discre

17、te domain, respectively,</p><p><b>　　(14)</b></p><p><b>　　(15)</b></p><p><b>　　where,</b></p><p><b>　　,</b></p>&

18、lt;p>　　Then, the observer equation adding an error correction term to is given by</p><p><b>　　(16)</b></p><p>　　Where K is the observer gain matrix and “＾ ” means the estimated qu

19、antity, and is the state variable estimated ahead one sampling period. Subtracting (15) from (16), the error dynamic equation of the observer is expressed as</p><p><b>　　(17)</b></p><p

20、>　　where . Here, it is assumed that the model parameters match well with the real ones. Fig. 3 shows the block diagram of the closed-loop state observer.</p><p>　　The state variable error depends only on

21、the initial error and is independent of the input. For (17) to converge to the zero state, the roots of the characteristic equation of (17) should be located within the unit circle. </p><p>　　Fig. 3. Closed-

22、loop state observer. </p><p>　　Fig. 4. Short pulse region. </p><p>　　4 EXPERIMENTS AND DISCUSSIONS</p><p>　　A. System Hardware Configuration</p><p>　　Fig. 5 shows the

23、 system hardware configuration. The source voltage is a three-phase, 110 [V].The input resistance and inductance are 0.06Ωand 3.3 mH, respectively. The dc link capacitance is 2350μF and the switching frequency of the PWM

24、 rectifier is 3.5 kHz.</p><p>　　Fig. 5. System hardware configuration.</p><p>　　Fig. 6. Dc link currents and corresponding phase currents (in sector V ).</p><p>　　The TMS320C31 DSP

25、chip operating at 33.3 MHz is used as a main processor and two 12-b A/D converters are used. One of them is dedicated for detecting the dc link current and the other is used for measuring the dc output voltage and the so

26、urce voltages and currents, where ac side quantities are just measured for performance comparison.</p><p>　　One of two internal timers in the DSP is employed to decide the PWM control period and the other is

27、 used to determine the dc link current interrupt. Considering the rectifier blanking time of 3.5 s, A/D conversion time of 2.6 s, and the other signal delay time, the minimum pulse width is set to 10 s.</p><p&

28、gt;　　Experimental Results </p><p>　　Fig. 6 shows measured dc link currents and phase currents. In case of sector V of the space vector diagram, the dc link current corresponds to for the switching state of

29、and for that of . Fig. 7(a) shows the raw dc link current before filtering. It has a lot of ringing components due to the resonance of the leakage inductance and the snubber capacitor. When the dc current is sampled at t

30、he end point of the active voltage vectors as shown in the figure, the measuring error can be reduced.</p><p>　　Fig. 7. Sampling of dc link currents.</p><p>　　Fig. 8. Estimated source voltage an

31、d current at starting. </p><p>　　To reduce this error further, the low pass filter should be employed, of which result is shown in Fig. 7(b). The cut-off frequency of the Butterworth’s second-order filter is

32、 112 kHz and its delay time is about 2 sec. Since the ringing frequency is 258 kHz and the switching frequency is 3.5 [kHz], the filtered signal without significant delay is acquired.</p><p>　　Fig. 8 shows t

33、he estimated source voltage and current at starting. With the proposed initial estimation strategy, the starting operation is well performed. Fig. 9 shows the phaseangle, magnitude, and waveform of the estimated source v

34、oltage, which coincide well with measured ones.</p><p>　　Fig. 10 shows the source voltage and current waveform at unity power factor. Figs. With the estimated quantities for the feedback control, the control

35、 performance is satisfactory. The dc voltage variation for load changes will be remarkably decreased if a feedforward control for theload current is added, which is possible without additional cur-rent sensor when the PW

36、M rectifier is combined with the PWM inverter for ac motor drives.</p><p>　　Fig. 9. Estimated source voltage in steady state.</p><p>　　(a) phase angle (b)magnitude (c) waveform.</p><p

37、>　　Fig. 10. Source voltage and current waveforms. </p><p>　　estimated (b) measured.</p><p>　　4 CONCLUSIONS</p><p>　　This paper proposed a novel control scheme of the PWM rectifie

38、rs without employing any ac input voltage and current sensors and with using dc voltage and current sensors only. Reducing the number of the sensors used decreases the system cost as well as improves the system reliabili

39、ty. The phase angle and the magnitude of the source voltage have been estimated by controlling the deviation between the rectifier current and its model current to be zero. For line current reconstruction, switching st&l

40、t;/p><p>　　無交流電動勢、電流傳感器的三相PWM整流器控制</p><p>　　Dong-Choon Lee, Member, IEEE, and Dae-Sik Lim</p><p>　　1 三相PWM 整流器</p><p><b>　　A 系統(tǒng)模型</b></p><p>　　圖一所

41、示為三相PWM整流器的主電路，電壓等式給出如下：</p><p><b>　　(1)</b></p><p>　　圖1 無交流傳感器三相PWM整流器</p><p>　　其中e，i和v分別是源電壓，線電流和整流器的輸入電壓，R和L分別是輸入電阻和輸入電感。當(dāng)已知線電壓峰值E，角頻率和初始相位角θ時，假定三相系統(tǒng)是平衡的，則源相位電壓可以表達(dá)為

42、</p><p><b>　　(2)</b></p><p><b>　　其中</b></p><p><b>　　(3)</b></p><p>　　一種基于估計相位角的變換矩陣，將三相變量變換成一個同步的，坐標(biāo)系，這個矩陣是</p><p><

43、b>　　(4)</b></p><p>　　將（1）式變?yōu)樽鴺?biāo)系使用式（4）</p><p><b>　　(5)</b></p><p>　　其中p是一個微分算子且</p><p>　　將（5）式寫成矢量形式</p><p><b>　　(6)</b><

44、;/p><p><b>　　其中</b></p><p>　　,,, (7)</p><p>　　用式（4）對（2）式進(jìn)行變換</p><p><b>　　(8)</b></p><p><b>　　其中</b>&

45、lt;/p><p><b>　　(9)</b></p><p>　　通過前向差分來接近微分的限幅，分別將（6）式和（8）式用離散域表示</p><p><b>　　(10)</b></p><p><b>　　(11)</b></p><p><b&g

46、t;　　其中，T是采樣周期</b></p><p>　　圖2 總的控制模塊圖</p><p><b>　　B 系統(tǒng)控制</b></p><p>　　PI控制器是用來調(diào)節(jié)直流輸出電壓和交流輸入電流的。對于解耦電流控制，交叉耦合項(xiàng)用前饋式補(bǔ)償，同時，源電壓作為擾動的補(bǔ)償。對于沒有過調(diào)的暫態(tài)響應(yīng)，引入anti-windup技術(shù)。消除源電壓

47、和線電流傳感器的總的控制模塊圖如圖2所示。源電壓和線電流的估計算法在以后的章節(jié)中介紹。</p><p><b>　　2預(yù)測電流估計</b></p><p>　　由于DSP存在計算時間，所以和不能立即計算。為了消除延遲的影響，可以使用狀態(tài)監(jiān)測器。另外，狀態(tài)監(jiān)測器可以對估計變量起到濾波作用。</p><p>　　將式（5）用狀態(tài)空間形式表達(dá)為<

48、;/p><p><b>　　(12)</b></p><p><b>　　(13)</b></p><p><b>　　其中</b></p><p><b>　　,,</b></p><p><b>　　,</b>

49、</p><p><b>　　Y是輸出。</b></p><p>　　分別將式（12）和式（13）分別變換成離散領(lǐng)域</p><p><b>　　(14)</b></p><p><b>　　(15)</b></p><p><b>　　其中&

50、lt;/b></p><p><b>　　,</b></p><p>　　則加入了誤差調(diào)整的監(jiān)測器等式為</p><p><b>　　(16)</b></p><p>　　其中，k是監(jiān)測器增益矩陣，“＾ ”是指估計量，是提前一個采樣周期估計的狀態(tài)變量。用式（15）和減去式（16），監(jiān)測器的動態(tài)

51、誤差等式表述為</p><p><b>　　(17)</b></p><p>　　其中這里，假設(shè)模型參數(shù)與真實(shí)系統(tǒng)吻合的很好。圖7所示是閉環(huán)狀態(tài)監(jiān)測器的模塊圖。</p><p>　　狀態(tài)變量誤差僅取決于初始誤差，與輸入無關(guān)。為了使式（17）趨于零狀態(tài)，典型等式（17）的根應(yīng)該限制在單位圓內(nèi)。</p><p>　　圖3

52、閉環(huán)狀態(tài)監(jiān)測器</p><p><b>　　圖4短脈沖區(qū)域</b></p><p><b>　　3實(shí)驗(yàn)與討論</b></p><p><b>　　A系統(tǒng)硬件構(gòu)造</b></p><p>　　圖5 系統(tǒng)硬件結(jié)構(gòu)</p><p>　　圖6 直流電流和相應(yīng)相

53、電流 (扇區(qū)5 ).</p><p>　　圖5所示是系統(tǒng)的硬件結(jié)構(gòu)圖。源電壓是三相110V。輸入電阻和電感分別為0.06Ω和3.3mH。直流側(cè)電容為2350μF，PWM整流器的開關(guān)切換頻率為3.5KHZ.使用TMS320C31 DSP芯片設(shè)定在33.3MHZ作為主處理器，同時用到兩個12位的A/D轉(zhuǎn)換器：一個用來檢測直流側(cè)電流，另一個用來檢測直流側(cè)輸出電壓、源電壓和電流。其中直流側(cè)數(shù)量只是為了性能比較而測量的。&

54、lt;/p><p>　　DSP內(nèi)部的兩個時鐘一個是用來決定PWM波的控制周期，另一個是用來決定直流側(cè)電流中斷?？紤]到整流器空白時間3.5μS，A/D轉(zhuǎn)換時間2.6μS和其他信號延遲時間，最小脈沖寬度設(shè)定為10μS.</p><p><b>　　C、實(shí)驗(yàn)結(jié)果</b></p><p>　　圖6所示是測得的直流側(cè)電流和相電流。假設(shè)空間矢量圖的扇區(qū)V，直流

55、側(cè)電流對應(yīng)于。圖7（a）所示是濾波之前未經(jīng)處理的直流側(cè)電流。因漏電感和緩沖電容的共振，會產(chǎn)生噪聲成分。如圖中所示，當(dāng)采樣動態(tài)電壓矢量末端的直流電流時，測量誤差可以減小。</p><p>　　圖7 直流側(cè)電流采樣</p><p>　　圖8 開始時的估計源電壓和電流 </p><p>　　為了進(jìn)一步減少誤差，可以使用低通濾波器，結(jié)果如圖7（b）所示。Butterwo

56、rth的第二順序?yàn)V波器的截止頻率是112KHZ，開關(guān)切換頻率為3.5KHZ，所以可以得到?jīng)]有顯著延遲的濾波信號。</p><p>　　圖8所示是開始時估計源電壓和電流。使用提出的初始估計策略，開始操作效果很好。圖9所示是估計源電壓的相位角、數(shù)值和波形。它們和測量的結(jié)果十分吻合。</p><p>　　圖10所是在單位功率因數(shù)時源電壓和電流波形。當(dāng)PWM整流器與逆變器相連時，在沒有額外電流傳感

57、器的情況下對交流汽車駕駛來說是可行的。</p><p>　　圖9穩(wěn)態(tài)時的估計源電壓. </p><p>　?。╝）相位角（b）數(shù)值 （c）波形</p><p>　　圖10 源電壓和電流波形</p><p>　?。╝）估計值 （b）測量值</p><p><b>　　4結(jié)論</b></p>

58、;<p>　　這篇文章提出了一種PWM整流器新穎的控制方法。這種方法沒有使用任何交流輸入電壓和電流傳感器，而僅僅使用直流電壓和電流傳感器。減少傳感器數(shù)量可以減少系統(tǒng)費(fèi)用的同時就提高系統(tǒng)的穩(wěn)定性。通過控制整流器的電流和它的模型電流的偏差為零，可以估計相位角和源電壓的數(shù)值。對于線電流重建，使用開關(guān)狀態(tài)和直流側(cè)電流測量。為了消除因微處理器計算時間所帶來的延遲影響，使用預(yù)測狀態(tài)監(jiān)測器。可以看出，估計算法對參數(shù)變化是健全的。整個算法