外文翻譯--三相電壓型pwm整流器建模和仿真研究

1、<p>　　重 慶 理 工 大 學</p><p>　　文 獻 翻 譯</p><p>　　二級學院 應用技術(shù)學院 </p><p>　　班 級 109217402 </p><p>　　學生姓名 康亮 學 號 10921740209</p><p>

2、　　譯 文 要 求</p><p>　　1、譯文內(nèi)容必須與課題（或?qū)I(yè)）內(nèi)容相關(guān)，并需注明詳細出處。</p><p>　　2、外文翻譯譯文不少于2000字；外文參考資料閱讀量至少3篇（相當于10萬外文字符以上）。</p><p>　　3、譯文原文（或復印件）應附在譯文后備查。</p><p>　　譯 文 評 閱</p>

3、<p>　　導師評語（應根據(jù)學?！白g文要求”，對學生外文翻譯的準確性、翻譯數(shù)量以及譯文的文字表述情況等作具體的評價）</p><p>　　指導教師： </p><p><b>　　年 月 日</b></p><p>　　三相電壓型PWM整流器建模和仿真研究</p><p>　　

4、摘要：三相電壓型PWM整流器（VSR）廣泛用于AC/DC/AC系統(tǒng)前端整流?？紤]到VSR本身非線性特點，建立適合于控制器設計上的數(shù)學模型比較，提出了一種狀態(tài)反饋解耦控制電流內(nèi)環(huán)和直流電壓平方外環(huán)的電壓型PWM整流器新型控制策略，基于功率平衡理論，采用解耦狀態(tài)反饋控制方法，分析并建立了三相電壓型PWM 整流器 d‐q 坐標系下的線性化數(shù)學模型。由于采用直流電壓平方外環(huán)，使典型的非線性模型線性化，控制器設計直觀精確，提高了直流電壓和網(wǎng)側(cè)電流

5、的跟蹤能力，改善了波形。提出了一種空間矢量的簡化算法，簡化了運算過程。在MATLAB/SIMULINK 環(huán)境中建立了仿真模型。仿真結(jié)果表明：所設計整流器具有優(yōu)良的穩(wěn)態(tài)性能和快速的動態(tài)響應，實現(xiàn)簡單，具有一定的實用價值。 </p><p>　　關(guān)鍵詞：電壓型PWM整流器；功率平衡；解耦狀態(tài)反饋；空間矢量脈寬調(diào)制；仿真</p><p><b>　　引言 </b></

6、p><p>　　在當今的電力系統(tǒng)當中大都采用二極管和相控轉(zhuǎn)換器。這種轉(zhuǎn)換器電路簡單，但缺點是線電流畸變嚴重和功率因數(shù)較低。為了解決這個問題，PWM 整流器的基于線電流波形整定的的各種功率因數(shù)校正技術(shù)被提出來了。</p><p>　　PWM 整流器有以下幾個優(yōu)勢比如:直流總線電壓的控制功率雙向流動單位功率因數(shù)、線電流正弦化。 </p><p>　　為了提高輸入功率因數(shù)和整

7、定輸入電流正弦化，整流裝置采用了許多控制技術(shù)，傳統(tǒng)的整流模型是多輸入多輸出非線性系統(tǒng)。整流器控制中最困難的就是非線性。</p><p>　　在優(yōu)秀的研究報告中，直接電流控制傳統(tǒng)的控制策略是建立功率因數(shù)補償?shù)膬?nèi)環(huán)和電壓調(diào)節(jié)的外環(huán)的雙環(huán)控制。大多數(shù)的系統(tǒng)參數(shù)依賴于PI調(diào)節(jié)器：輸出電壓控制環(huán)會產(chǎn)生電流內(nèi)環(huán)的參考電流的參考指數(shù)或振幅。 電流內(nèi)環(huán)的作用是是三相交流負載的電流跟隨給定信號的變化。</p><

8、;p>　　本文著重探討了VSR的建模和控制。以一種新的基于電力電量平衡方程來取代原有的非線性方程。然后應用非線性輸入變換使改進后的模型線性的。提出了一種簡化算法空間矢量PWM整流器。該算法避免了傳統(tǒng)方法的查表的正弦或反三角和復雜計算的需要是直接計算空間電壓矢量的責任周期跟蹤參考電壓矢量在每一個環(huán)節(jié)上的空間矢量。</p><p>　　1、 VSR的建模和控制</p><p>　　1.1

9、VSR在dq坐標系下的數(shù)學模型</p><p>　　三相電壓型電路的主電路如圖 1 所示，每個半導體開關(guān)由一個 IGBT 和并行的二極管組成。這里 ua，ub，uc分別為三相平衡電壓源的相電壓，ia，ib，ic為相電流，vdc是直流輸出電壓，R和L分別代表濾波電抗器的電阻和電感，C是平滑電容，RL是直流側(cè)負載，il 是負載電流。 </p><p>　　以下公式描述了整流器在 dq 坐標下的

10、動態(tài)特性：</p><p>　　在這里urd=Sdvdc, urq=Sqvdc,urd,urq,和Sd,Sq分別是整流器輸入電壓，在同步旋轉(zhuǎn)dq坐標系的開關(guān)函數(shù)。ud,uq和id,iq分別為同步旋轉(zhuǎn)dq坐標下的電壓源和電流,ω為角頻率。 </p><p>　　圖 1，三相電壓型整流器主電路</p><p>　　1.2 電流環(huán)狀態(tài)反饋解耦方法</p>&

11、lt;p>　　在上述的非線性方程中，公式（1）（2）說明sd,sq與狀態(tài)變量vdc有關(guān)，urd=Sdvdc和urq=Sqvdc,說明urd和Sd ,urq和Sq沒有動態(tài)關(guān)系，因此一個非線性輸入變換可以用于修改將舊的輸入變量 sd,sq變成urd ,urq,而且模型說明d‐q電流和耦合電壓wliq和wlid有關(guān)系，而且受主電壓ud,uq以及urd和urq的影響。公式（1）和(2）中的urd和urq表示為公式（4）（5）。</p

12、><p>　　將公式（4）（5）帶入公式（1）（2），被控變量和新輸入的最關(guān)系是線性和解耦的非線性表達，VSR 的預期關(guān)系是：</p><p>　　從等式中我們可以看到，兩個軸的電流是完全解耦的，與只和期望的id與iq是有關(guān)系的，電壓環(huán)和電流環(huán)采用簡單的PI控制方法。</p><p>　　1.3 外環(huán)電壓設計</p><p>　　公式（3）描述了

13、Vdc的模型，功率平衡方程可以用來輔助替代方程模型。吸收的有功功率交流電流功率（Pac）和有功功率轉(zhuǎn)換器直流功率（Pdc）表達:</p><p>　　Pac和Pdc的關(guān)系是：</p><p>　　Pac=Pdc+Ploss (10)</p><p>　　Ploss包括電阻R功率損耗以及開關(guān)和VSR傳導損失，電阻R通常很小，它實際上是合理的忽視它的能量損失

14、，整流器損失是比電阻R損失功率大，但它們?nèi)匀皇强偣β屎苄〉囊徊糠郑虼?，忽略整流器損失沒有明顯的損失整流器準確性。如果更精確的表示損失，需要整流器可以表示一個小電阻RL,直流側(cè)總電阻用RL表示，從Pac=Pdc中可以看出，下面是動態(tài)結(jié)果：</p><p><b>　　重新整流公式得：</b></p><p>　　由Vdc的單相特性，以為變量，公式（12）就會變成線性的

15、，將公式（8）帶入公式（12）得到</p><p>　　這是的動態(tài)方程和輸出的狀態(tài)變量，是輸入，設計一個簡單的PI控制器能夠調(diào)節(jié)直流電壓無穩(wěn)態(tài)誤差，ud是可測量的，實際的輸出變量id從中得到，電流內(nèi)環(huán)的結(jié)果為id的參考值。</p><p>　　圖2顯示內(nèi)部電流回路與狀態(tài)反饋解耦和VSR外環(huán)控制系統(tǒng)。</p><p>　　圖2，三相VSR的雙閉環(huán)控制模塊</p&

16、gt;<p>　　2.空間電壓矢量合成</p><p>　　當?shù)玫絬rd和urq后，通過dq變換到αβ變換得到精確的直流電壓命令和直流總線電壓。</p><p>　　根據(jù)圖1開關(guān)狀態(tài)的橋式整流電路，橋式整流器電壓可以假設8個狀態(tài)電壓矢量（V0到V7）。V1到V7是六個確定的非零矢量，V0和V7是圖3中所示的兩個零矢量。三相輸入電壓分為六個60°，如圖4所示：<

17、/p><p><b>　　定義</b></p><p>　　圖3 PWM橋式整流器α-β變換空間矢量表示</p><p>　　圖4 三相輸入電壓六個分區(qū)</p><p>　　N=sign(B0)+2sign(B1)=4sign(B2) (16)</p><p>　　在圖5所示，信號

18、分為6個60°間隔，相對于另一個信號的跡象，它滿足了那個標志兩個信號幅值都是一樣的。在每個分區(qū)，并沒有明顯變化。設置的值，每個都是獨一無二的。例如，再間隔1，B0是正的，B1，B2是負的。</p><p>　　圖5，B0，B1，B2六個分區(qū)</p><p>　　其中的矢量是基于表達式（6）的，如圖4表示，矢量與N的一致關(guān)系如表1所示。</p><p>&l

19、t;b>　　表一</b></p><p>　　三相電壓可視為一個電壓矢量對。有許多不同的方法合成，根據(jù)調(diào)制的不同組合八個向量。這些方法，可以使兩相調(diào)制的開關(guān)損耗減少，在一個工作循環(huán)內(nèi)其中一個開關(guān)應該總是開或關(guān)。理想的參考矢量是在每一個子環(huán)平均取樣時間Ts和實現(xiàn)了三個最近的空間向量的平均向量。例如，在圖3中所示的參考矢量，電壓Vs和角度θ和電流I用矢量1，矢量2和零矢量表示。三個持續(xù)的空間向量T1

20、、T2、TZ分別計算為：</p><p>　　其他矢量合成與矢量合成方法是相似的，通用的變量X,Y,Z的通用矢量表達如下：</p><p>　　對于任何參考向量，持續(xù)兩個時間空間向量，如列表2。</p><p><b>　　3、仿真</b></p><p>　　基于前面的分析，利用圖1的三相VSR的MATLAB/SIMU

21、LINK仿真，利用IGBT的實驗負載和以下參數(shù)：</p><p>　　uRMS≈220V,L ≈3mH,R≈0.1Ω,C ≈4700Mf,RL≈16Ω,vdc=700V.</p><p>　　下面的兩個數(shù)據(jù)總結(jié)simulation的仿真結(jié)果。圖6的結(jié)果顯示了瞬態(tài)響應輸出電壓，第二個數(shù)字顯示輸入電流的瞬態(tài)響應。在回路負載RL=16Ω時仿真開始時刻直流母線電壓停留在二極管整流器的水平。然后，應

22、用控制負載電阻和輸出電壓增加到預期直流電壓值。圖7顯示所需的電壓和電流在同一側(cè)。我們能看到電流與電壓同相位。</p><p>　　圖6，直流電壓動態(tài)仿真結(jié)果</p><p><b>　　4、總結(jié)</b></p><p>　　本文中，給出了一個非線性變換方法推導三相VSR。一種新的控制策略是應用前面介紹的狀態(tài)反饋解耦的電流內(nèi)環(huán)和本文介紹的基于狀態(tài)

23、空間解耦的電壓外環(huán)，利用非線性輸入的轉(zhuǎn)變，傳統(tǒng)的非線性模型可以變換為線性模型。這一改善使設計的控制器變得簡單明了。介紹了SVPWM算法描述和驗證。仿真結(jié)果表明它具有更好的控制精度，更少的開關(guān)動作、計算簡便、容易實現(xiàn)，更好的利用直流電壓。</p><p>　　圖7，A相電壓、電流仿真結(jié)果</p><p><b>　　文獻原文</b></p><p&

24、gt;　　Modeling and Simulation Research for Three-Phase Voltage Source PWM Rectifier</p><p>　　Abstract:Pulse-Width Modulated three-phase Voltage Source Rectifier(VSR)is the building blocks of the most of AC/DC

25、/AC systems as the front-end rectifier. The major difficulty in control is caused by the nonlinearities in the rectifier model. The linear mathematical model of VSR in d-q coordinates was deduced with analysis based on t

26、he power balance equation. A new control strategy using inner current loop with state feedback decoupling and outer voltage square loop was proposed.Nonlinear input </p><p>　　Keywords:VSR;power balance equat

27、ion;state feedback decoupling; SVPWM;simulation </p><p>　　Introduction </p><p>　　Diode and phase-controlled converters constitute the largest segment of power electronics that interface to the e

28、lectric utility system today. These converter circuits are simple but the disadvantages are large distortion in line current and poor power factor. To combat these problems the PWM rectifier various power factor correcti

29、on(PFC) techniques based on active wave shaping of the line current have been proposed.</p><p>　　The PWM rectifier offers several advantages such as: control of DC bus voltage,bi-directional power flow unity

30、 power factor and sinusoidal line current. </p><p>　　Many control techniques have been adopted for these rectification devices to improve the input power factor and shape the input current of the rectifier i

31、nto sinusoidal waveform. In actual implementations the direct current control scheme is widely adopted. The conventional rectifier model is a multi-input multi-output nonlinear system. The difficulty in controlling the r

32、ectifiers is mainly due to the nonlinearity.</p><p>　　As reported in the excellent survey traditional control strategies in the direct current control scheme establish two loops: a line current inner loop fo

33、r power factor compensation and an output voltage outer loop for voltage regulation. The most uses system parameters dependent Proportional Integral (PI) regulator: for the output voltage control loop which can generate

34、the modulation index or the amplitude of the reference current for the inner PWM input current control loops. The main task of </p><p>　　This paper focuses on the modeling and control of the VSR.A new equati

35、on based on power balance is introduced to replace the original nonlinear equation. Then,nonlinear input transformation is applied to make the improved model linear. A simplified algorithm is proposed for space vector PW

36、M rectifier. This algorithm avoids the look-up tables of sine or arc-tangent and complex calculations needed in the conventional methods by directly calculating the duty cycles of space voltage vectors which tr</p>

37、<p>　　1 Modeling and Control of VSR </p><p>　　1.1 The mathematical model of VSR in d-q coordinates</p><p>　　The main circuit diagram of the three-phase voltage source rectifier structure

38、is shown in Fig.1.Each power semiconductor switch consists of an IGBT connected in parallel with a diode. Where ua ,ub and uc are the phase voltages of three phase balanced voltage source and ia ,ib and ic are phase curr

39、ents vdc is the DC output voltage R and L mean resistance and inductance of filter reactor respectively C is smoothing capacitor across the dc bus RL is the DC side load and iL is load current. </p><p>　　The

40、 following equations describe the dynamical behavior of the boost type rectifier in Park coordinated or in d-q:</p><p>　　Where,urd=Sdvdc, urq=Sqvdc,urd,urq,and Sd,Sq are input voltage of rectifier,switch fun

41、ction in synchronous rotating d-q coordinate respectively. Ud,uq and id,iq are voltage source current in synchronous rotating d-q coordinate respectively. ω is angular frequency.</p><p>　　Fig.1 Circuit sche

42、matic of three-phase two-level boost-type rectifier</p><p>　　1.2 Decoupled state-feedback control method of current loop</p><p>　　In the above nonlinear model equation 1 and equation2 show that

43、both input variables Sd and Sq are coupled with the state variable vdc. The fact that urd Sd vdc and urq=Sq vdc, shows that there is no dynamics between urd and Sd or urq and Sq.Therefore a nonlinear input transformation

44、 can be used to modify the old input variables Sd and Sq to the new input variables urd and urq. Moreover the model shows that d-q current is related with both coupling voltages ωLiq and ωLid, and main voltages ud an<

45、/p><p>　　Putting equation (4)and(5)into equation(1)and(2)the nonlinear expression is such that the final relation between the controlled variables and the new inputs is linear and decoupled. Thus,the expected r

46、elations in the VSR are,</p><p>　　We can see from equation that the two axis current are totally decoupled. urd’ and urq, are only related with id and iq respectively.The simple proportional-integral(PI)cont

47、rollers are adopted in the current and voltage regulation.</p><p>　　1.3 Design of outer voltage square loop</p><p>　　Equation (3) describes the dynamics of vdc. Power balance equation can be use

48、d to derive an alternate equation for vdc dynamics. The active power absorbed from the ac source(Pac) and the active power delivered to the converter dc-side (Pdc) are expressed by: </p><p>　　The relationshi

49、p between Pac and Pdc is: </p><p>　　Pac=Pdc+Ploss (10)</p><p>　　Where Ploss includes the power loss in the resistor R as well as the switching and conduction losses in the VSR. The resist

50、ance R is always very small and thus it is practically reasonable to neglect its power loss. The rectifier losses are larger than the power loss in R but still they count for a small portion of the total power. Therefore

51、,the rectifier losses can also be neglected without noticeable loss of accuracy. If better accuracy is desired the rectifier losses can be represented by a </p><p>　　Which can be rearranged in following form

52、: </p><p>　　Due to uni-directional nature of vdc ,Taking vdo2 as the variable,(12) will become linear.</p><p>　　Putting equation (8) into equation(12),</p><p>　　This is a first-orde

53、r dynamic equation with vdc2 as the state variable as well as the output,and Pac as the input. A simple PI controller can be designed to regulate the DC voltage with no steady state error. Since ud is measurable,the actu

54、al input variable id can be derived from Pac. The result is actually the reference value of id for the current inner control loop.</p><p>　　Fig.2 displays inner current loop with state feedback decoupling an

55、d outer voltage square loop control system for VSR.</p><p>　　Fig.2 Block diagram of double close-loop control for three-phase VSR</p><p>　　2 Voltage Space-vector Synthesization</p><p

56、>　　When the urd and urq acquired the SVPWM method is realized through d-q to α-βtransformation to trace the AC current command exactly and regulate the DC bus voltage.</p><p>　　Depending on the switching

57、state on the circuit Fig.1 the bridge rectifier leg voltages can assume 8 possible distinct states represented as voltage vectors (V0 to V7). V1 to V6 are six fixed nonzero vectorsV0 and V7 are two zero vectors as shown

58、in the Fig.3.The input three phase voltage are divided into six 60°input intervals,as shown in Fig.4. </p><p>　　Defining: </p><p>　　Fig.3α-β space vector representation of the PWM bridge r

59、ectifier leg voltage</p><p>　　Fig.4 Six intervals of input three voltage</p><p>　　N=sign(B0)+2sign(B1)=4sign(B2) (16)</p><p>　　As shown in Fig.5the signals are divided i

60、nto six 60° intervals it satisfies that the signs of the amplitudes of two signals are the same and opposite to the sign of another signal. And no sign change occurs during each interval. The value of N in every sec

61、tor is unique.In interval 1,for example,B0 is positive,B1 and B2 are negative.</p><p>　　Fig.5 Six intervals of B0 B1 and B2</p><p>　　The sector in which is depends on the expression (6) Compar

62、ed with Fig.4,it is obvious that the corresponding relations between value N and sector are seen in Table1.</p><p>　　Table 1 Determination of sector of based on N</p><p>　　Three-phase voltage c

63、an be treated as a voltage vector Vs. There are many different methods of modulation to synthesize according to the different combinations eight vectors. Among these methods,the two-phase modulation can make switching l

64、oss minimize,in which one switch should be always set ON or OFF in one working cycle.The desired reference vector is sampled in every sub-cycle Ts and realized by time averaging the three nearest space vectors in the spa

65、ce vector plane. For example the refere</p><p>　　The vector synthetic method of other sector is similar. The expressions which is developed on the universal variable X,Y,Z are shown as following: </p>

66、<p>　　For any reference vector the duration time of two space vectors are assigned as Table2.</p><p>　　3 Simulation Results</p><p>　　Based on the former analysis the MATLAB/SIMULINK simula

67、tion model for the VSR of Fig.1with the test load was implemented using IGBT modules and following values:</p><p>　　uRMS≈220V,L ≈3mH,R≈0.1Ω,C ≈4700Mf,RL≈16Ω,vdc=700V.</p><p>　　The following two

68、figures summarize the results of the simulation.Fig.6 shows the transient response of the output voltage. The second figure shows transient response of input current. In this simulation at start time the dc bus voltage r

69、ests at the diode rectifier level with a resistive load of RL=16Ω.Then,the control action is applied keeping the load resistance and the output voltage increases to the desired dc value.Fig.7 shows the voltage and curre

70、nt on line side. We can see the current of</p><p>　　Fig.6 Simulation result for DC-Link voltage dynamics</p><p>　　4 conclusion </p><p>　　In this paper a no linear transformation is

71、used to derive for three-phase VSR.A new control strategy using inner current loop with state feedback decoupling and outer voltage square loop based on the space vector modulation is introduced in this paper.By using no

72、nlinear input transformation the conventional nonlinear models can be improved to linear models. This improvement makes the design of the controller become straightforward. A novel SVPWM algorithm has been described and